Ultrasonic diagnostic systems are known to the art for providing real-time, cross-sectional (tomographic) images of the human body. Ultrasonic energy is focused at a target tissue by a transmit beamformer, and ultrasonic energy modulated by the target tissue is focused by a receive beamformer. In one class of modern ultrasonic diagnostic systems, the transmit and receive beamformers are both coupled to a single time-shared transducer array, and the receive beamformer focuses energy reflected by the target tissue. Alternately, the receive beamformer may be responsive to transmitted energy which has passed through the target tissue. The receive beamformer may provide images, color Doppler or spectral Doppler information regarding the target tissue, or combinations thereof. Typically, the transmit beamformer controls delays with which electrical signals are applied to individual transducers to achieve constructive interference of ultrasonic energy at the target tissue. Similarly, the receive beamformer applies carefully controlled delays to the transducer signals prior to summing to steer and focus the receive beam as desired.
Typically, a receive beamformer must provide both a large delay range and a high delay resolution. The delay resolution will set the focus accuracy of the system, and is much less than one cycle of the carrier frequency. Delay resolution accuracy of one 16th of a carrier cycle is good by current standards, but accuracy of one 64th of a cycle will afford improved image resolution. The delay range needed is set by the maximum path length difference between outer transducers in extreme cases of focusing or steering. This delay range can be as high as 45 carrier cycles (or more), for instance with a 128-element transducer array steered at 45.degree. and having a transducer spacing of one half the carrier wavelength. The ratio of range to resolution accuracy can therefore be 1000:1 or even 2000:1. For straight-forward implementations, this corresponds to an analog delay line with 1000 or 2000 taps, or a digital RAM of that size. Furthermore, such a digital RAM must operate at high speeds; for a carrier cycle resolution of 1/32 with a carrier of 5 MHz, a 160 MHz operating speed would be needed, and even higher speeds would be needed either for a higher frequency carrier or for greater delay resolution.
To compound these difficulties, the delay profile (the relative delays of the different channels) must be dynamically changed during the receive interval if tracking focus is to be maintained at a point on the outwardly propagating transmit wavefront.
For these reasons, much effort has been directed at reducing the complexity of the receive hardware used to perform the delay for focusing and steering. Maslak, U.S. Pat. No. 4,140,022, recognized that the delay applied to each receive signal can be divided into a coarse delay and a fine delay, with the latter implemented as phase shifting. This division is based on the fact that there are two parts to the receive signal: the carrier and the envelope. The envelope contains information about the target along the range direction, and the envelope has much reduced delay resolution requirements as compared with the carrier. The carrier, specifically the relative phasing of the carrier across the channels, contains information about the location of the target in the azimuth direction. The carrier delay requires very fine resolution, but this can be achieved through phase rotation rather than delays. Maslak '022 uses heterodyning with a variable phase clock to achieve this phase rotation.
This division of the total delay into coarse and fine delays significantly simplifies the hardware required for receive beamforming. First, by phase shifting the signals on a per-channel basis (for fine delay), a greatly reduced number of taps are required on the analog delay lines (for coarse delay). Furthermore, adjacent channels can be summed after fine delay and before coarse delay, thereby reducing the number of delay lines required.
The basic approach set out in Maslak '022 (dividing the total delay into a coarse delay and a fine delay) has since been implemented in many forms.
Maslak, et al., U.S. Pat. No. 4,550,607, and Maslak, et al., U.S. Pat. No. 4,699,009, describe a simplified, cyclically symmetrical summing delay line architecture to further leverage the separation of coarse and fine delay, as well as a dynamic focusing method in which the settings of the delay lines for the coarse delay need not be changed during active receive. Dynamic focus is performed by adjusting the phase of the heterodyne reference signals for each channel. Again, after the per-channel application of fine delay, pairs of adjacent channels are summed prior to coarse delay.
Riley, et al., U.S. Pat. No. 4,662,223, increases to four the number of channels summed together after the per-channel application of a first delay, and prior to a second delay means. Riley '223 also notes that the output frequency of the mixers for the first delay could be at baseband as well as IF. Riley '223 teaches the use of analog delay lines for the second delay means, and in practical applications such analog delay lines are generally configured for coarse, not fine delays.
Saugeon, et al., U.S. Pat. No. 4,829,491, takes a more direct approach to the fine delay: each channel includes a predelay element which has only about one carrier cycle of total delay, but includes many taps for fine delay resolution. This predelay element is followed by a coarse delay structure. Four adjacent channels are summed after the individual application of fine delay, prior to the coarse delay means. Again, it is the fine delay means that are updated for dynamic focusing.
Saugeon '491 implements the coarse delay means with digital hardware, and notes that separation of total delay into fine and coarse delays allows lower clock frequencies. In a digital memory-based delay implementation, the clock period sets the delay resolution. Saugeon '491 also teaches a digital implementation of the fine delay using a variable-length shift register operating at a high clock frequency. The division into fine and coarse delays in this case limits the length of the high-frequency shift register, and the summation of four adjacent channels reduces the size, power, and cost of the coarse delay means.
Another early beamformer system implemented with digital delays is described in O'Donnell, et al., U.S. Pat. Nos. 4,809,184 and 4,839,652. O'Donnell '184 notes that delay resolution needed for accurate beamforming requires very high frequency operations for a simple RAM implementation--a much higher frequency than needed for Nyquist sampling (digitization) of the received signal. O'Donnell '184 addresses this problem by making fine adjustments to the sample clock for the analog-to-digital converter for each channel, and thereby allowing the use of coarse digital delays.
Larson, U.S. Pat. No. 5,263,004, teaches another method in which a surface-acoustic-wave (SAW) device is used as a fine delay element. The delay resolution of this technique is very good, but the SAW device limits the total delay range, so this device is followed by a coarse delay means.
Fife, et al., U.S. Pat. No. 5,269,307, describes an ultrasound imaging system in which fine delay phasing is performed using a vector modulator. A fine phase shift is created by splitting the received signal into two components with 90.degree. phase separation by means of an all-pass-filter network. These two components are then multiplied with sin.PHI. and cos.PHI. weights to implement a phase shift .PHI.. This approach allows nearly continuous phase resolution when compared to earlier heterodyne techniques. Pairs of the outputs of these fine delay phase shifters are summed together prior to a coarse delay means, which can be implemented using delay lines or digital circuitry.
Matsushima, et al., U.S. Pat. No. 5,375,470, describes fine delay phase shifters that are implemented in a similar way, but at a different position in the signal processing path. Whereas Fife '307 teaches a 90.degree. phase separation, followed by sin.PHI. or cos.PHI. weighting and summing of adjacent pairs, Matsushima '470 teaches as vector weighting followed by summing of adjacent pairs, and finally the 90.degree. phase separation. By applying the sin.PHI. and cos.PHI. weighting before the 90.degree. separation, the channels can be combined at an earlier stage to allow further reductions in hardware size, power, and cost. In both Matsushima '470 and Fife '307, the 90.degree. phase shift is implemented with a delay line of one-quarter of the carrier frequency.
The system described by Matsushima '470 also includes dynamic focusing in which both the fine delay and the coarse delay are dynamically updated. Because changes to the coarse delay are, by nature, very large and would therefore tend to cause artifacts in the image, a large portion of the focusing apparatus is duplicated, and the two are time-interleaved with smoothed transitions between them. Advantageously, a common delay means is used, which includes both the coarse delays and the delays for the 90.degree. shifts, with the coarse settings chosen by preceding switch matrixes. The settings for one switch matrix are updated when the other matrix is active to reduce image artifacts.
In systems having a fine/coarse separation of delay, the optimal coarse delay settings can be computed knowing only the target location and then rounding. However, a compromise must be made when paired channels require different optimal coarse delays. Furthermore, the coarse delays are generally not updated for dynamic focusing, or are updated at great hardware expense, as Matsushima '470 illustrates. Both the combined channel coarse setting compromise and a lack of coarse delay dynamic updates has limited the performance of prior art receive beamformers with fine/coarse delay separation.
Also, in all systems utilizing a fine/coarse separation of delay, the fine delay control must be generated knowing both the target location and the specifics of the coarse delay settings. Increased beamforming accuracy requires more frequent and more precise control of focusing and steering delays, making the generation of control signals a more significant task. Furthermore, the wide variety of transducer geometries and scan formats now proliferating also makes generation of the control signals a more significant task.
In the systems described above, the initial delay has been implemented using analog techniques, though often only digital timing signals (clocks) need to be involved (i.e Maslak '022 and O'Donnell '184).
With the advance of digital integrated circuit technology, the fine delay has, predictably, been implemented digitally. O'Donnell et al., U.S. Pat. No. 4,983,970, uses a CORDIC multiplier for phase rotation of a digitized ultrasound signal. A conventional digital memory (FIFO) is used for the coarse delay, with the phasing of the multiplier providing fine delay. In fact, the FIFO precedes the CORDIC multiplier, reversing the order of fine and coarse delays.
Lipschutz, U.S. Pat. No. 5,345,426, describes an alternative digital fine delay mechanism based on an FIR filter interpolator. In Lipschutz the coarse delay also precedes the fine delay.
Because it is possible to implement the fine and coarse delay in each of these digital systems in the same digital hardware, the difficulty of generating the control signals for the fine delay means is greatly eased.
For instance, in the digital beamformer described in co-pending U.S. patent application Ser. No. 08/432,615, assigned to the assignee of the present invention, there is coarse delay provided by a memory, fine delay provided by a complex demodulator for phase shift, and a control structure in which numerous delay adjustment terms are computed and combined using more precision than available in the coarse delay memory. The resulting least significant bits are rolled into the fine delay control.
FIG. 29 shows the architecture of these digital beamformers at a high level. Each receive signal is associated with a respective signal processing channel, and each channel includes an analog pre-conditioning circuit (APC), an analog to digital converter (ADC), and a digital beamformer processor (DBP). The outputs of the DBP's are summed in a summer.
The digital beamforming process can perform the summation en masse as shown, or incrementally (one channel at a time), but the process must include both very fine per channel delay adjustment and beamforming summation. Typically it also includes per-channel amplitude weighting (apodization) and demodulation to base-band. The fine delay adjustment is typically implemented as a phase rotation of an analytic baseband signal.
The analog pre-conditioning in this case can include signal compression (DGC/TGC), low-noise amplification, and other manipulations that are identically done to every channel.
The system of FIG. 29 requires an ADC and a DBP for every channel, and the result is a system of considerable size and cost, with high power requirements. This is so despite the fact that the signals received by adjacent channels may be largely the same, requiring similar firmware controls for the digital beamformer processors.
These last three examples are all-digital in that all the beamforming steps are performed digitally. There are significant advantages to all-digital beamformers. The accuracy of the delay and phasing is exceptional, and unlike analog implementations, there will be little or no drift over the operating life of the machine. This accuracy improvement is particularly marked for dynamic receive focusing, where the local hardware can interpolate the controls to give nearly perfect focus for all output samples. It is also easier to include many advanced features with digital beamformers, such as adaptive phase aberration correction, multiple receive beamformation, system autocalibration and the like.
However, the cost and power required per channel in a digital beamformer are substantial. Whereas the analog beamformer of Maslak '607 required only one ADC per beamformer, all-digital beamformers require one ADC per channel. Also, the delay and phasing digital hardware, though highly integrated, is expensive and requires substantial power.
Furthermore, improved imaging performance is possible with the use of two-dimensional transducer arrays, but at the expense of the concomitant need for more channels.
The present invention addresses the need for an ultrasonic beamformer with performance heretofore associated only with all-digital beamformers, but with reduced cost, power, and size.